PWM capacitor control

ABSTRACT

Methods, systems, and devices for controlling a variable capacitor. One aspect features a variable capacitance device that includes a capacitor, a first transistor, a second transistor, and control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including detecting a zero-crossing of an input current at a first time. Switching off the first transistor. Estimating a first delay period for switching the first transistor on when a voltage across the capacitor is zero. Switching on the first transistor after the first delay period from the first time. Detecting a zero-crossing of the input current at a second time. Switching off the second transistor. Estimating a second delay period for switching the second transistor on when a voltage across the capacitor is zero. Switching on the second transistor after the second delay period from the second time.

CROSS-REFERENCE TO RELATED APPLICATION

Under 35 U.S.C. § 120, this application is a divisional of and claims priority to U.S. patent application Ser. No. 16/038,569, filed on Jul. 18, 2018, which is a continuation of and claims priority to U.S. patent application Ser. No. 15/427,186, filed on Feb. 8, 2017, now U.S. Pat. No. 10,063,104, which claims priority to U.S. Provisional Patent Application Nos. 62/292,474, filed on Feb. 8, 2016; 62/376,217, filed on Aug. 17, 2016; 62/407,010, filed on Oct. 12, 2016; and 62/408,204 filed on Oct. 14, 2016. The entire contents of each of these priority applications are incorporated herein by reference in their entirety.

BACKGROUND

Power electronics may rely on electronic circuits such as rectifiers, AC (Alternating Current) to DC (Direct Current) converters, impedance matching circuits, and other power electronics to condition, monitor, maintain, and/or modify the characteristics of the voltage and/or current used to provide power to electronic devices. Circuit components with adjustable impedance can used in such contexts to modify the voltage and/or current characteristics of various electronic devices. Controlling such components to avoid damage can be challenging. Moreover, present adjustable impedance circuit components may sacrifice efficiency power losses in order to ensure safe operation. For example, PWM controlled reactive components (e.g., capacitors and inductors) may rely on lossy diode conduction currents to clamp component voltages at zero while transistors are switched in order to avoid damaging current surges through the transistors.

SUMMARY

In general, the disclosure features control systems and processes for controlling a variable reactive circuit component, such as a PWM controlled capacitor. The devices and process described herein can be used in a variety of contexts, including impedance matching networks, implantable devices, cell phone and other mobile computing device chargers, and chargers for electric vehicles.

In a first aspect, the disclosure features a variable capacitance device that includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including detecting a first zero-crossing of an input current at a first time. Switching off the first transistor after a first delay period from the first time. A length of the first delay period can be controlled by an input value. Detecting a second zero-crossing of the input current at a second time, after the first time. Measuring an elapsed time between switching off the first transistor and detecting the second zero-crossing. Setting a counter based on the elapsed time. Switching on the first transistor after a second delay period based on the counter.

In a second aspect, the disclosure features a high-voltage impedance matching system that includes an impedance matching network and a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including detecting a first zero-crossing of an input current at a first time. Switching off the first transistor after a first delay period from the first time. A length of the first delay period can be controlled by an input value. Detecting a second zero-crossing of the input current at a second time, after the first time. Measuring an elapsed time between switching off the first transistor and detecting the second zero-crossing. Setting a counter based on the elapsed time. Switching on the first transistor after a second delay period based on the counter.

In a third aspect, the disclosure features a wireless energy transfer system that includes an inductive coil electrically connected to a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including detecting a first zero-crossing of an input current at a first time. Switching off the first transistor after a first delay period from the first time. A length of the first delay period can be controlled by an input value. Detecting a second zero-crossing of the input current at a second time, after the first time. Measuring an elapsed time between switching off the first transistor and detecting the second zero-crossing. Setting a counter based on the elapsed time. Switching on the first transistor after a second delay period based on the counter.

These and the following aspects can each optionally include one or more of the following features.

In some implementations, the operations of the control circuitry include switching off the second transistor after the first delay period from the second time. Detecting a third zero-crossing of the input current at a third time, after the second time. Measuring a second elapsed time between switching off the second transistor and detecting the third zero-crossing. Setting a second counter based on the second elapsed time. Switching on the second transistor after a third delay period based on the second counter.

In some implementations, the effective capacitance of the capacitor is controlled by the input value.

In some implementations, the input value is a phase delay value, and the first delay period is equal to φ/360° T, where φ represents the phase delay value and T represents a period of the input current.

In some implementations, setting the counter based on the elapsed time includes setting the counter to the measured elapsed time plus a predetermined delay time.

In some implementations, the predetermined time delay less than 800 ns.

In some implementations, the first and second transistors are silicon MOSFET transistors, silicon carbide MOSFET transistors, or gallium nitride MOSFET transistors.

In some implementations, switching on the first transistor includes switching on the first transistor in response to detecting body-diode conduction through the first transistor.

In some implementations, the body-diode conduction through the first transistor indicates a zero voltage condition across the capacitor.

In a fourth aspect, the disclosure features a variable capacitance device that includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including determining a first delay period based on a phase delay value. Determining a second delay period based on the phase delay value, where the second delay period being longer than the first delay period. Detecting a first zero-crossing of an input current at a first time. Switching off the first transistor after the first delay period from the first time. Switching on the first transistor after the second delay period from the first time. Detecting a second zero-crossing of the input current at a second time, after the first time. Switching off the second transistor after the first delay period from the second time. Switching on the second transistor after the second delay period from the second time.

In a fifth aspect, the disclosure features a high-voltage impedance matching system that includes an impedance matching network and a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including determining a first delay period based on a phase delay value. Determining a second delay period based on the phase delay value, where the second delay period being longer than the first delay period. Detecting a first zero-crossing of an input current at a first time. Switching off the first transistor after the first delay period from the first time. Switching on the first transistor after the second delay period from the first time. Detecting a second zero-crossing of the input current at a second time, after the first time. Switching off the second transistor after the first delay period from the second time. Switching on the second transistor after the second delay period from the second time.

In a sixth aspect, the disclosure features a wireless energy transfer system that includes an inductive coil electrically connected to a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including determining a first delay period based on a phase delay value. Determining a second delay period based on the phase delay value, where the second delay period being longer than the first delay period. Detecting a first zero-crossing of an input current at a first time. Switching off the first transistor after the first delay period from the first time. Switching on the first transistor after the second delay period from the first time. Detecting a second zero-crossing of the input current at a second time, after the first time. Switching off the second transistor after the first delay period from the second time. Switching on the second transistor after the second delay period from the second time.

These and the other aspects can each optionally include one or more of the following features.

In some implementations, the effective capacitance of the capacitor is controlled by the phase delay value.

In some implementations, the first delay period is equal to

${\frac{\varphi}{360{^\circ}}T},$ where φ represents the phase delay value and T represents a period of the input current.

In some implementations, the second delay period is equal to

${\frac{{360{^\circ}} - \varphi}{360{^\circ}}T},$ where φ represents the phase delay value and T represents a period of the input current.

In some implementations, switching on the first transistor after the second delay period from the first time includes switching on the first transistor following a fixed time delay after the second delay period from the first time.

In some implementations, switching on the first transistor after the second delay period from the first time includes switching on the first transistor in response to detecting body-diode conduction through the first transistor.

In some implementations, the body-diode conduction through the first transistor indicates a zero voltage condition across the capacitor.

In some implementations, the first and second transistors are silicon MOSFET transistors, silicon carbide MOSFET transistors, or gallium nitride MOSFET transistors.

In a seventh aspect, the disclosure features a variable capacitance device that includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including generating an alternating ramp signal having peaks and troughs that are timed to correspond with zero-crossings of an input current. Switching off the first transistor in response to the ramp signal crossing a first reference value. Switching on the first transistor after the ramp signal crosses the first reference value and in response to detecting body-diode conduction through the first transistor. Switching off the second transistor in response to the ramp signal crossing a second reference value. Switching on the second transistor after the ramp signal crosses the second reference value and in response to detecting body-diode conduction through the first transistor.

In an eighth aspect, the disclosure features a high-voltage impedance matching system that includes an impedance matching network and a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including generating an alternating ramp signal having peaks and troughs that are timed to correspond with zero-crossings of an input current. Switching off the first transistor in response to the ramp signal crossing a first reference value. Switching on the first transistor after the ramp signal crosses the first reference value and in response to detecting body-diode conduction through the first transistor. Switching off the second transistor in response to the ramp signal crossing a second reference value. Switching on the second transistor after the ramp signal crosses the second reference value and in response to detecting body-diode conduction through the first transistor.

In a ninth aspect, the disclosure features a wireless energy transfer system that includes an inductive coil electrically connected to a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including generating an alternating ramp signal having peaks and troughs that are timed to correspond with zero-crossings of an input current. Switching off the first transistor in response to the ramp signal crossing a first reference value. Switching on the first transistor after the ramp signal crosses the first reference value and in response to detecting body-diode conduction through the first transistor. Switching off the second transistor in response to the ramp signal crossing a second reference value. Switching on the second transistor after the ramp signal crosses the second reference value and in response to detecting body-diode conduction through the first transistor.

These and the other aspects can each optionally include one or more of the following features.

In some implementations, the effective capacitance of the capacitor is controlled by the first and second reference values.

In some implementations, the second reference value has a value that is the negative of the first reference value.

In some implementations, switching on the first transistor includes switching on the first transistor following a fixed time delay after the ramp signal crosses the first reference value following the peak in the ramp signal.

In some implementations, switching on the first transistor includes switching on the first transistor after the ramp signal crosses the first reference value following a peak in the ramp signal and in response to detecting body-diode conduction through the first transistor.

In some implementations, the body-diode conduction through the first transistor indicates a zero voltage condition across the capacitor.

In some implementations, the first and second transistors are silicon MOSFET transistors, silicon carbide MOSFET transistors, or gallium nitride MOSFET transistors.

In a tenth aspect, the disclosure features a variable capacitance device that includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including detecting a zero-crossing of an input current at a first time. Switching off the first transistor. Estimating, based on an input value, a first delay period for switching the first transistor on when a voltage across the capacitor is zero. Switching on the first transistor after the first delay period from the first time. Detecting a zero-crossing of the input current at a second time. Switching off the second transistor. Estimating, based on the input value, a second delay period for switching the second transistor on when a voltage across the capacitor is zero. Switching on the second transistor after the second delay period from the second time.

In an eleventh aspect, the disclosure features a high-voltage impedance matching system that includes an impedance matching network and a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including detecting a zero-crossing of an input current at a first time. Switching off the first transistor. Estimating, based on an input value, a first delay period for switching the first transistor on when a voltage across the capacitor is zero. Switching on the first transistor after the first delay period from the first time. Detecting a zero-crossing of the input current at a second time. Switching off the second transistor. Estimating, based on the input value, a second delay period for switching the second transistor on when a voltage across the capacitor is zero. Switching on the second transistor after the second delay period from the second time.

In a twelfth aspect, the disclosure features a wireless energy transfer system that includes an inductive coil electrically connected to a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including detecting a zero-crossing of an input current at a first time. Switching off the first transistor. Estimating, based on an input value, a first delay period for switching the first transistor on when a voltage across the capacitor is zero. Switching on the first transistor after the first delay period from the first time. Detecting a zero-crossing of the input current at a second time. Switching off the second transistor. Estimating, based on the input value, a second delay period for switching the second transistor on when a voltage across the capacitor is zero. Switching on the second transistor after the second delay period from the second time.

These and the other aspects can each optionally include one or more of the following features.

In some implementations, the effective capacitance of the capacitor is controlled by the input value.

In some implementations, the first delay period is equal to

${\frac{{360{^\circ}} - \varphi}{360{^\circ}}T},$ where φ represents the input value and T represents a period of the input current.

In some implementations, switching on the first transistor after the first delay period from the first time includes switching on the first transistor following a fixed time delay after the first delay period from the first time.

In some implementations, switching on the first transistor after the first delay period from the first time includes switching on the first transistor in response to detecting body-diode conduction through the first transistor.

In some implementations, the body-diode conduction through the first transistor indicates a zero voltage condition across the capacitor.

In some implementations, the first and second transistors are silicon MOSFET transistors, silicon carbide MOSFET transistors, or gallium nitride MOSFET transistors.

In some implementations, the operations of the control circuitry include determining a third delay period, based on the input value, and switching off the first transistor includes switching off the first transistor after the third delay period from the first time.

In some implementations, the third delay period is equal to

${\frac{\varphi}{360{^\circ}}T},$ where φ represents the input value and T represents a period of the input current.

In some implementations, the operations of the control circuitry include determining a fourth delay period, based on the input value, and switching off the second transistor includes switching off the second transistor after the fourth delay period from the second time.

In some implementations, the fourth delay period is equal to

${\frac{\varphi}{360{^\circ}}T},$ where φ represents the input value and T represents a period of the input current.

In a thirteenth aspect, the disclosure features a variable capacitance device that includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including switching off the first transistor at a first time. Switching on the first transistor after detecting a current through a first diode associated with the first transistor. Switching off the second transistor at a second time. Switching on the second transistor after detecting a current through a second diode associated with the second transistor.

In a fourteenth aspect, the disclosure features a high-voltage impedance matching system that includes an impedance matching network and a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including switching off the first transistor at a first time. Switching on the first transistor after detecting a current through a first diode associated with the first transistor. Switching off the second transistor at a second time. Switching on the second transistor after detecting a current through a second diode associated with the second transistor.

In a fifteenth aspect, the disclosure features a wireless energy transfer system that includes an inductive coil electrically connected to a variable capacitance device. The variable capacitance device includes a capacitor, a first transistor, a second transistor, and control circuitry. The first transistor includes a first-transistor source terminal, a first-transistor drain terminal, and a first-transistor gate terminal. The first-transistor drain terminal is electrically connected to a first terminal of the capacitor. The first-transistor gate terminal is coupled to the control circuitry. The second transistor includes a second-transistor source terminal, a second-transistor drain terminal, and a second-transistor a gate terminal. The second-transistor drain terminal is electrically connected to a second terminal of the capacitor. The second-transistor source terminal is electrically connected to the second-transistor source terminal. The second-transistor gate terminal is coupled to the control circuitry. The control circuitry is configured to adjust an effective capacitance of the capacitor by performing operations including switching off the first transistor at a first time. Switching on the first transistor after detecting a current through a first diode associated with the first transistor. Switching off the second transistor at a second time. Switching on the second transistor after detecting a current through a second diode associated with the second transistor.

These and the other aspects can each optionally include one or more of the following features.

In some implementations, the first diode is electrically connected in parallel with the first transistor, and the second diode is electrically connected in parallel with the second transistor.

In some implementations, the first diode is a body-diode of the first transistor, and the second diode is a body-diode of the second transistor.

Some implementations include a body diode conduction sensor electrically connected to the first transistor and the second transistor.

In some implementations, the body diode conduction sensor is coupled to the control circuitry and provides signals indicating a start of body diode conduction through the first diode and through the second diode.

In some implementations, the body diode conduction sensor includes a sense resistor electrically connected between the first transistor and the second transistor.

In some implementations, the body diode conduction sensor includes an operational amplifier comprising a first input terminal electrically connected to a one terminal of the sense resistor and a second input terminal electrically connected to another terminal of the sense resistor.

In some implementations, the body diode conduction sensor is configured to operate using a bipolar voltage supply.

In some implementations, the body diode conduction sensor is configured to operate using a unipolar voltage supply.

In some implementations, the first and second transistors are silicon MOSFET transistors, silicon carbide MOSFET transistors, or gallium nitride MOSFET transistors.

In a sixteenth aspect, the disclosure features an impedance matching network of a wireless power transmission system that includes first and second transistor switching elements having internal body diodes or external antiparallel diodes associated therewith. A PWM-switched capacitor coupled across the first and second switching elements. A controller coupled to control the first and second switching elements to minimize the body diode conduction time by steering current flow away from body diodes into the channels of the first and second transistor switching elements. This and the other aspects can each optionally include one or more of the following features.

In some implementations, the controller includes zero voltage switching ZVS circuitry to control switching to occur when a voltage across the PWM-switched capacitor and the first and second switching elements is near or at zero.

In some implementations, the controller is a mixed signal implementation.

In some implementations, the controller is a digital signal implementation and includes a microcontroller, a zero-crossing detection stage having an output sent to the microcontroller, and a power stage to which the zero-crossing detection stage is coupled. The the zero-crossing detection stage includes a comparator and a current sensor (908) that produces a voltage signal for the comparator. The power stage includes gate drivers for driving the first and second transistor switching elements and signal isolation for input signals to the gate drivers generated by the microcontroller.

In some implementations, the controller is a digital signal implementation that includes starting a cycle of a switching period; detecting a zero-crossing of an input current by a zero-crossing detector when the input current is rising; scheduling the first transistor switching element to turn off at time t₂ where t₂=φ/360°·T and T is a period of the input current and phase φ sets an equivalent capacitance of the PWM-switched capacitor to approximately

${C_{eq} = {C{1 \cdot \frac{1}{2 - {\left( {{2\varphi} - {\sin 2\varphi}} \right)/\pi^{\prime}}}}}};$ scheduling the second transistor switching element to turn on at a time t₅, where

$t_{5} = {{\frac{{360^{\circ}} - \varphi}{360^{\circ}} \cdot T} + T_{delay}}$ and delay T_(delay) is adjusted so zero-voltage switching is ensured for all operating conditions; finishing the cycle by turning on the second transistor switching element M2; turning off the first transistor switching element; detecting zero-crossing of the input current when the input current is falling; scheduling the second transistor switching element to turn off at time t₆, where t₆=T/2+φ/360°·T; scheduling the second transistor switching element to turn on at time t₉, where

${t_{9} = {{\frac{{480{^\circ}} - \varphi}{360{^\circ}} \cdot T} + T_{del}}};$ zero voltage switching first transistor switching element; turning on the first transistor switching element; turning off the second transistor switching element; detecting zero-crossing of the input current to start a next cycle when the input current is rising; scheduling switching element to turn off after t=φ/360°·T; zero voltage switching the second transistor switching element; turning on the second transistor switching element; transitioning to a start of a next cycle.

In some implementations, the first and second transistor switching elements are MOSFET devices.

In some implementations, the first and second transistor switching elements are galium nitride (GaN) or silicon carbide (SiC) transistor switching elements.

In some implementations, the controller is a gate control module for providing a first gate control signal for the first switching element and a second gate control signal for the second switching element, as well as a reference potential for a node between the gates of the first and second switching elements.

In some implementations, the PWM-switched capacitor provides an equivalent capacitance of

${Ceq} = {C1\frac{1}{2 - {\left( {{2\varphi} - {\sin 2\varphi}} \right)/\pi}}}$ where C1 is an impedance value of the capacitor and φ is a phase delay.

In a seventeenth aspect, the disclosure features a wireless power transmission system that includes a source-side circuit and a device-side circuit. The source-side circuit includes an inverter for powering the source-side circuit, the impedance matching network the of any of the above described aspects, and a source resonator. The device-side circuit includes a device resonator a device impedance matching network, and a rectifier. The impedance matching network couples, with a coupling factor, oscillating electromagnetic energy to the device-side circuit where the oscillating electromagnetic energy is converted by the rectifier.

In some implementations, the source-side circuit includes a source resonator coil, a series capacitor, a parallel capacitor, a capacitor, and an inductor, where the capacitor is the PWM-switched capacitor.

Particular implementations of the subject matter described in this specification can be implemented so as to realize one or more of the following advantages. Implementations may reduce body-diode (or antiparallel diode) conduction times associated with power losses in switching transistors, and thereby, improve operational efficiency and/or thermal management. Implementations may permit the use of a wider array of transistors, including those having relative large forward body-diode voltage drops, for example, gallium nitride (GaN) of silicon carbide (SiC) transistors. Implementations may provide improved tolerance of input currents that have harmonic content, such as a triangular waveform, a trapezoidal waveform, a square waveform, or a waveform with sinusoidal characteristics with significant harmonic content.

Embodiments of the devices, circuits, and systems disclosed can also include any of the other features disclosed herein, including features disclosed in combination with different embodiments, and in any combination as appropriate.

The details of one or more implementations of the subject matter described in this specification are set forth in the accompanying drawings and the description below. Other features, aspects, and advantages of the subject matter will be apparent from the description, the drawings, and the claims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic representation of a wireless energy transfer system.

FIG. 2 is a schematic circuit representation of wireless energy transfer system including an illustrative impedance matching network(IMN) having one or more tunable capacitors.

FIG. 3A-3B shows schematic representations of a PWM capacitor.

FIG. 4 is a diagrammatic representation of mixed signal implementation of the control of a PWM capacitor.

FIG. 5A is a diagrammatic representation of a modulator of the mixed signal implementation of FIG. 4 .

FIG. 5B is a graphical representation showing waveforms associated with the modulator of FIG. 5A.

FIG. 6A is a diagrammatic representation of a pulse shaping circuitry of the mixed signal implementation of FIG. 4 . FIG. 6B is a graphical representation showing waveforms associated with the modulator of FIG. 6A.

FIG. 7A is a diagrammatic representation of a power stage of the mixed signal implementation of FIG. 4 . FIG. 7B is a graphical representation showing waveforms associated with the modulator of FIG. 6A. FIG. 7C is a zoomed in view of the graphical representation shown in FIG. 7B.

FIGS. 8A-8F are graphical representations of measured waveforms associated with a mixed signal implementation of the control of a PWM capacitor.

FIG. 9 is a diagrammatic representation of a digital implementation of the control of a PWM capacitor.

FIG. 10A is a flowchart of an exemplary process for the control of a PWM capacitor.

FIG. 10B is a timing diagram of process described in FIGS. 10A and 10C.

FIG. 10C is a flowchart of another exemplary process for the control of a PWM capacitor.

FIGS. 11A-11F are graphical representations of measured waveforms associated with a digital implementation of the control of a PWM capacitor.

FIG. 12 is a schematic representation of a PWM capacitor switching system.

FIG. 13A is an example circuit implementation of a peak detector that can form a part of the system of FIG. 12 .

FIG. 13B is a waveform diagram showing illustrative waveforms for the circuit of FIG. 13 .

FIG. 13C is another example circuit implementation of a peak detector that can form a part of the system of FIG. 12

FIGS. 14A and 14B are example circuit implementation of current shape analysis that can form a part of the system of FIG. 12

FIG. 14C is a waveform diagram showing illustrative waveforms for the circuits of FIGS. 14A and 14B

FIG. 15A is an example circuit implementation of an over current protection circuitry that can form a part of the system of FIG. 12 .

FIG. 15B is a waveform diagram showing illustrative waveforms for the circuit of FIG. 15A.

FIG. 16A is an example circuit implementation of an incremental over current protection circuitry that can form a part of the system of FIG. 12 .

FIG. 16B is a waveform diagram showing illustrative waveforms for the circuit of FIG. 16A.

FIG. 17A is an example circuit implementation of an over voltage protection circuitry that can form a part of the system of FIG. 12 .

FIG. 17B is a waveform diagram showing illustrative waveforms for the circuit of FIG. 17A.

FIG. 18 is an example circuit implementation of a zero-crossing detector that can form a part of the system of FIG. 12 .

FIG. 19 is an example circuit implementation of a bandpass filter/integrator circuitry to generate a ramp signal that can form a part of the system of FIG. 12 .

FIG. 20 is an example circuit implementation of a PWM signal generator that can form a part of the system of FIG. 12 .

FIG. 21 is a schematic representation of a PWM capacitor switching system.

FIG. 22 is a schematic representation of a PWM capacitor switching system having ZVS.

FIG. 23A is an example circuit implementation of a zero-crossing detector.

FIG. 23B is an example circuit implementation of a body diode conduction sensor.

FIGS. 24A-24E are waveform diagrams showing illustrative waveforms for the circuit of FIG. 22 .

FIGS. 25A-25C are waveform diagrams showing illustrative waveforms for the circuits of FIGS. 22 and 23 .

FIG. 26 is an example circuit implementation of the modulator of FIG. 22 .

FIGS. 27A-27E are waveform diagrams showing illustrative waveforms for the circuits of FIG. 22 and FIG. 26 .

FIG. 28A is an example circuit implementation of a signal delay circuit and FIG. 28B is an example circuit implementation of a signal conditioning circuit.

FIGS. 29A-29D are waveform diagrams showing illustrative waveforms for the circuits of FIG. 22 and FIGS. 28A and 28B.

FIGS. 30A-30F are waveform diagrams showing illustrative waveforms for the circuits of FIG. 22 and FIGS. 28A and 28B.

FIGS. 31A and 31B show example waveforms for a circuit shown in FIG. 31C with silicon MOSFETs without automatic ZVS and example waveforms for the circuit shown in FIG. 31C with automatic ZVS.

FIG. 32 shows example waveforms for a circuit with silicon carbide MOSFETs without and with automatic ZVS.

FIG. 33 shows example thermal imaging of a circuit without and with automatic ZVS.

FIG. 34 shows a schematic representation of an illustrative computer that can perform at least a portion of the processing described herein.

Like reference numbers and designations in the various drawings indicate like elements.

DETAILED DESCRIPTION

In general, the disclosure features control systems and processes for controlling a variable reactive circuit component. Implementations of the present disclosure are described in the context of a circuit including a PWM-switched capacitor coupled across first and second switching elements (e.g., transistors). Implementations disclosed herein may minimize diode conduction time for external antiparallel or internal body diodes associated with the first and second switching elements. Implementations of the PWM-switched capacitor circuit can operate with sinusoidal input currents containing significantly higher harmonic content than conventional circuits. Shorting a PWM-switched capacitor when a zero voltage is not present can be undesirable and may damage the switching elements and/or increase power loss. Implementations discussed herein control the first and second switching elements to minimize the body diode conduction time (dead time) by steering current flow away from body diodes into the transistor (e.g. MOSFET) channel. In doing so, losses due to diode voltage drops are minimized. Accordingly, implementations may provide efficient circuit operation while maintaining zero voltage switching. Implementations can be implemented with a computer processor, microcontroller, digital-signal processor, FPGA, CPLD, or any other programmable processing device to generate gate control signals, in mixed signal configurations, and in digital circuitry. Furthermore, implementations of the present disclosure provide variable capacitor control that allow for efficient operation over the entire range of conditions encountered by impedance matching networks in highly-resonant wireless power transfer systems (HRWPT) system such as high-power vehicle charging systems, for example.

Control of the PWM capacitor can be implemented in several ways, such as in a mixed signal (analog and digital) implementation and/or a digital signal implementation. These implementations are described more fully below. Advantages of the disclosed implementations include the following:

In some implementations, the body-diode (or antiparallel diode) conduction time can be adjustable and significantly reduced. Such reductions in body-diode (or antiparallel diode) conduction time reduces MOSFET losses and improves efficiency and thermal management of power electronics.

In some implementations, the PWM capacitor control techniques permit the use of a wider array of transistors, including those having relative large forward body-diode voltage drops, for example, gallium nitride (GaN) of silicon carbide (SiC) transistors.

In some implementations, the PWM capacitor provides improved tolerance of input currents that have harmonic content, such as a triangular waveform, a trapezoidal waveform, a square waveform, or a waveform with sinusoidal characteristics with significant harmonic content. This is an advantage over conventional control methods that may require purely sinusoidal currents. For example, to achieve a purely sinusoidal current, filtering components can be added to the circuit, adding cost and component count. In some implementations, the PWM capacitor can tolerate transients, such as at the start-up of an associated system.

FIG. 1 shows a high level functional block diagram of an exemplary implementation of a wireless power transfer system 100 having PWM switched capacitors. Input power to the system can be provided by wall power (AC mains), for example, which is converted to DC in an AC/DC converter block 102. In some implementations, a DC voltage can be provided directly from a battery or other DC supply. In some implementations, the AC/DC converter block 102 may include a power factor correction (PFC) stage. The PFC, in addition to converting the AC input (for example, at 50 or 60 Hz) to DC, can condition the current such that the current is substantially in phase with the voltage.

A switching inverter 104 converts the DC voltage into AC voltage waveform (e.g., a high-frequency AC voltage waveform). The AC voltage waveform outputted by the inverter 104 is used to drive a source resonator 106. In some implementations, the frequency of the AC voltage waveform may be in the range of 80 to 90 kHz. In some implementations, the frequency of the AC voltage waveform may be in the range of 1 kHz to 15 MHz. In some implementations, the inverter 104 includes an amplifier.

A source impedance matching network (IMN) 108 couples the inverter 104 output to the source resonator 106. The source IMN 108 can enable efficient switching-amplifier operation. For example, class D or E switching amplifiers are suitable in many applications and can require an inductive load impedance for highest efficiency. The source IMN 108 can transform effective impedances of the source resonator as seen by the inverter 104. The source resonator impedance can be, for example, loaded by being electromagnetically coupled to a device resonator 110 and/or output load. For example, the magnetic field generated by the source resonator 106 couples to the device resonator 110, thereby inducing a corresponding voltage. This energy is coupled out of the device resonator 110 to, for example, directly power a load or charge a battery.

A device impedance matching network (IMN) 112 can be used to efficiently couple energy from the device resonator 110 to a load 114 and optimize power transfer between source resonator 106 and device resonator 110. Device IMN 112 can transform the impedance of a load 114 into an effective load impedance seen by the device resonator 110 which more closely matches the source impedance to increase system efficiency. For loads requiring a DC voltage, a rectifier 116 converts the received AC power into DC. In some implementations, the source 118 and device 120 a further include filters, sensors, and other components.

The impedance matching networks (IMNs) 108, 112 can be designed to maximize the power delivered to the load 114 at a desired frequency (e.g., 80-90 kHz, 100-200 kHz, 6.78 MHz) or to improve power transfer efficiency. The impedance matching components in the IMNs 108, 112 can be chosen and connected so as to preserve a high-quality factor (Q) value of resonators 106, 110. Depending on the operating conditions, the components in the IMNs 108, 112 can be tuned to control the power delivered for the power supply to the load 114, for example improve efficient wireless transfer of power.

The IMNs (108, 112) can have components including, but not limited to, a capacitor or networks of capacitors, an inductor or networks of inductors, or various combinations of capacitors, inductors, diodes, switches, and resistors. The components of the IMNs can be adjustable and/or variable and can be controlled to affect the efficiency and operating point of the system. Impedance matching can be performed by varying capacitance, varying inductance, controlling the connection point of the resonator, adjusting the permeability of a magnetic material, controlling a bias field, adjusting the frequency of excitation, and the like. The impedance matching can use or include any number or combination of varactors, varactor arrays, switched elements, capacitor banks, switched and tunable elements, reverse bias diodes, air gap capacitors, compression capacitors, barium zirconium titanate (BZT) electrically tuned capacitors, microelectromechanical systems (MEMS)-tunable capacitors, voltage variable dielectrics, transformer coupled tuning circuits, and the like. The variable components can be mechanically tuned, thermally tuned, electrically tuned, piezo-electrically tuned, and the like. Elements of the impedance matching can be silicon devices, gallium nitride devices, silicon carbide devices, and the like. The elements can be chosen to withstand high currents, high voltages, high powers, or any combination of current, voltage, and power. The elements can be chosen to be high-Q elements.

Control circuitry in a source 118 and/or device 120 monitors impedance differences between the source 118 and the device 120 and provides control signals to tune respective IMNs 108, 112 or components thereof. In some implementations, the IMNs 108, 112 can include a fixed IMN and a dynamic IMN. For example, a fixed IMN may provide impedance matching between portions of the system with static impedances or to grossly tune a circuit to a known dynamic impedance range. In some implementations, a dynamic IMN can be further composed of a coarsely adjustable components and/or finely adjustable components. For example, the coarsely adjustable components can permit coarse impedance adjustments within a dynamic impedance range whereas the finely adjustable components can be used to fine tune the overall impedance of the IMN(s). In another example, the coarsely adjustable components can attain impedance matching within a desirable impedance range and the finely adjustable components can achieve a more precise impedance around a target within the desirable impedance range.

FIG. 2 shows an exemplary embodiment of a wireless power transmission system 200 having an inverter 202 powering source-side circuit (which includes source resonator and source IMN) 204, which couples, with coupling factor k, oscillating electromagnetic energy to the device-side circuit (which includes device resonator and device IMN) 206. This oscillating energy is then converted by the rectifier 208. The source-side circuit 204 components include source resonator coil L_(s) 210, series capacitor C_(1s) 212 (in position 1), parallel capacitor C_(2s) 214 (in position 2), and capacitor C_(3s) 216 and inductor L_(3s) 218 (in position 3). In the illustrative embodiment, capacitor C_(3s) 216 can include one or more variable capacitors. For example, the variable capacitor can be a pulse width modulation (PWM) controlled capacitor. Note the each of the components listed may represent networks or groups of components and that components in at least position 1 and 3 can be balance. The device-side circuit 206 components can include device resonator coil L_(d) 222, series capacitor C_(1d) 224 (in position 1), parallel capacitor C_(2d) 226 (in position 2), and capacitor C_(3d) 228 and inductor L_(3d) 230 (in position 3). The capacitor C_(3d) 228 can be include one or more variable capacitors, such as a PWM capacitor. The PWM switched capacitors 216, 228 can promote efficient wireless energy transfer, as described more fully below.

IMNs 108 and 112 can have a wide range of circuit implementations with various components having impedances to meet the needs of a particular application. For example, U.S. Pat. No. 8,461,719 to Kesler et al., which is incorporated herein by reference in its entirety, discloses a variety of tunable impedance network configurations, such as in FIGS. 28 a-37 b . In some implementations, each of the components shown in FIG. 2 may represent networks or groups of components. In addition, while illustrative embodiments are shown and described in conjunction with highly resonant wireless energy transfer systems, implementations of PWM switched components described herein are applicable to a wide range of applications in which it is desirable to achieve a given equivalent impedance and minimize diode conduction times.

FIG. 3A shows an illustrative circuit implementation of a PWM-switched capacitor C1. In some implementations, an equivalent capacitance can be determined as

${{Ceq} = {C1\frac{1}{2 - {\left( {{2\varphi} - {\sin 2\varphi}} \right)/\pi}}}},$ where C1 is an impedance value of the capacitor and φ is an input phase delay, as described more fully below.

First and second switching elements M1, M2 are coupled back-to-back across or in parallel to capacitor C1. The first and second switching elements M1, M2 can be MOSFET devices. A gate control circuitry 300 provides a first gate control signal g1 for the first switching element M1 and a second gate control signal g2 for the second switching element M2. In some implementations, gate control circuitry 300 provides a reference potential s12 for a node between the gates of the first and second switching elements M1, M2.

Input current I₁ flows into a first node N1 and current I_(C1) flows out of the first node to capacitor C1. Current I₂ flows out of the first node N1 into the drain terminal of the first switching element M1. The capacitor C1 is coupled between the V_(cap+) and V_(cap−) nodes to define the voltage across the capacitor. In some implementations, the circuit can include a first sensor S1 to sense MOSFET body diode conduction and a second sensor S2 to sense current through the switched capacitor, as described more fully below. In some implementations, the switching elements M1, M2 may be silicon MOSFETs. FIG. 3B shows the circuit of FIG. 3A with external diodes D1, D2 positioned in antiparallel configuration relative to M1, M2. These diodes D1, D2 can be external diodes or the body diodes of switching elements M1, M2, as such the term “body-diodes” is used herein to refer collectively to both a power transistor body-diode or an external antiparallel diode associated with a transistor as shown in FIGS. 3A and 3B. The switching elements can include, but are not limited to silicon transistors, silicon carbide transistors, gallium nitride transistors, MOSFET (metal oxide semiconductor field-effect transistors), IGBT (insulated-gate bipolar transistors), JFET (junction gate field-effect transistor), or BJT (bipolar junction transistors).

Mixed-Signal Implementation

FIG. 4 shows a diagram of an exemplary embodiment of a mixed-signal implementation of the control of a PWM capacitor. This implementation includes a controller 400 in communication with a controller interface 402, which is in communication with modulator 404. The modulator 404 communicates with pulse shaping circuit 406 for zero voltage switching (ZVS) control. The pulse shaping circuit 406 communicates with power stage 408, which communicates with the modulator 404. These blocks are described further below.

FIG. 5A shows a diagram of an exemplary embodiment of the controller interface 402 and modulator 404 of FIG. 4 . The modulator stage can include reference signal generation, current sensor output, zero-crossing detection, ramp generation, and PWM generation. A microcontroller (μC) sets control signal V_(r) which is used to control the equivalent capacitance of the PWM capacitor. Control signal V_(r) can be a DC voltage signal or pulse-width of modulated signal with average voltage V_(ref). Reference signal generator 502 creates V_(ref+) and V_(ref−) voltages that have approximately the same absolute value but opposing sign. The output of current sensor 504 is provided to the zero-crossing detector 506. The output of the current sensor 504 is a generally sinusoidal signal that represents input current I₁ to the PWM capacitor. In some implementations, I₁ can have significant harmonic content. Zero-crossing detector 506 detects zero-crossings of the current I₁.

Zero-crossing detector 506 outputs a square-wave signal V_(zc)=V_(zc−)−V_(zc+). In other words, the output of the zero-crossing detector 506 can be, for example, a signal with +5V amplitude when I₁ is negative and −5V amplitude when I₁ is positive. Ramp generator 508 converts square-wave signal V_(zc) to a ramp signal V_(ramp) using, for example, an integrator circuit. Ramp generator 508 provides a ramp signal that a positive slope when the current I₁ is positive and a negative slope when the current I₁ is negative. In addition, the peaks of the ramp signal may correspond to zero-crossings of current I₁, as shown in subplot III of FIG. 5B.

High-frequency filter 510, composed of C20 and R49, eliminates any DC bias that may exist at the output of operational amplifier U2. PWM generation 512 creates switching functions PWM_M1 and PWM_M2 that control the switching elements M1 and M2. Two comparators 514 a, 514 b are used to produce these signals from V_(ramp), V_(ref+), and V_(ref−).

FIG. 5B shows plots of waveforms of modulator 404 as described in FIG. 5A. Subplot I shows current measurement I(L1) at current sense transformer L1 in the power stage 408, further described below. Note that this current is not purely sinusoidal and has some harmonic content. In some embodiments, the current may be stepped down using a transformer (as indicated by L1:L2 in FIG. 5A) with a ratio of 1:100 (or similar), so that the current can be handled by the components in the modulator circuit. Subplot II shows a voltage measurement V(V_(zc−), V_(zc+)) between nodes V_(zc−) and V_(zc+) at the zero-crossing detector 506. Subplot III shows voltage measurement V(Vramp), having a triangular waveform, at the output of the ramp generator 508. Subplot IV shows voltage measurement V(PWM_M1), in a dashed line, at the output of the PWM generation comparator 514 a and V(PWM_M2), in a solid line, at the output of the PWM generation comparator 514 b. Subplot V shows voltage waveform V_(c1) of a voltage measurement between nodes V_(cap+) and V_(cap−) and thus, the effective capacitance measured between nodes V_(cap+) and V_(cap−). This effective capacitance includes the contributions of capacitance C1 and switching elements M1 and M2. Line 516 shows that, in some implementations, the rising edge of switching element M1 turn-on signal has to be delayed for ZVS operation of switching element M1.

FIG. 6A shows a diagram of an exemplary embodiment of pulse shaping circuitry 406 for ZVS control of FIG. 4 . The pulse shaping circuitry 406 includes subcircuit 602 with output PWM1 and subcircuit 604 with output PWM2. In some implementations, inputs PWM_M1 and PWM_M2 may not be used to directly drive switching elements M1 and M2 due to a possible non-zero voltage condition at turn-on on capacitor C1. Thus, signals PWM_M1 and PWM_M2 may be conditioned by subcircuits 602 and 604 to create desirable signals PWM1 and PWM2, respectively, which are then used to drive switching elements, M1, M2. In some implementations, subcircuits 602, 604 act as multiplexers with selection signals en0 to en3.

For example, turning on switching elements M1, M2 at non-zero voltage of capacitor C1 may lead to excessive losses, physical damage to switching elements, or both. Pulse shaping circuit 406 can condition signals PWM_M1 and PWM_M2 by delaying turn-on edge of PWM_M1 and PWM_M2 such that zero-voltage turn-on of M1 and M2 can be achieved. Manually adjustable pulse shaping circuit can be configured adjust the ZVS condition on-the-fly for different input currents I₁. Note that ZVS can be manually adjustable by activating any of the selection signals en0 to en3. The body diode of a MOSFET is on before ZVS turn-on. The conduction time of body-diode is greatly reduced from conventional operation but it is not minimal. As shown, pulse shaping circuit 406 is implemented using logic gates, however, in some implementations, a digital multiplexer circuit can also be used to achieve similar results.

FIG. 6B shows plots of waveforms of pulse shaping circuitry 406 as described in FIG. 6A. Subplot I shows current measurement I(L1) at L1 of the current transformer. The current sense transformer includes L1 (at the power stage 408) and L2 (at the modulator 404). Subpolt II shows voltage measurement V(PWM_M1), in a dashed line, at the input of subcircuit 602 and V(PWM_M2), in a solid line, at the input of subcircuit 604. Subplot III shows voltage waveforms of voltage measured V(g1,s12) between gate control signal g1 and reference potential s12 in a dashed line and voltage measured V(g2,s12) between gate control signal g1 and reference potential s12 in a solid line. Subplot IV shows voltage waveform V_(C1) of a voltage measurement between nodes V_(cap+) and V_(cap−) and thus, the effective capacitance measured between nodes V_(cap+) and V_(cap−). Window 606 shows the delay in the turn-on of M1 such that ZVS is achieved for I₁ currents that differ from a purely sinusoidal signal.

FIG. 7A shows a diagram of an exemplary embodiment of power stage 408 of FIG. 4 . The power stage 408 contains capacitor C1, back-to-back switching element pair M1 and M2, current sensor (current sense transformer) L1 that measures the current through PWM capacitor (I₁), gate drivers 702 that drive M1 and M2, isolated power supply 704 for gate drivers, signal isolation 706 for gate driver input signal. The input signals are generated by the modulator 404 and pulse shaping 406 stages. In some implementations, the current sense signal form L1 is supplied to modulator 404.

FIG. 7B shows plots of waveforms of power stage 408 as described in FIG. 7A. Subplot I shows voltage waveforms of voltage simulated V(g1,s12) between gate control signal g1 and reference potential s12 in a dashed line and voltage measured V(g2,s12) between gate control signal g2 and reference potential s12 in a solid line. Voltage waveforms V(g1,s12) and V(g2,s12) overlap in amplitude but are shifted by 180 degrees or a half of the switching period relative to one another such that the positive half cycle of V(C1) is symmetrical to negative half cycle of V(C1). Subplot II shows a current waveform I(L1) at the current sense transformer L1 (see power stage 408 in FIG. 5C). This current is not purely sinusoidal and has some harmonic content. Subplot III shows a current waveform of I₂ that flows out of the first node N1 into the drain terminal of the first switching element M1. Subplot W shows a current waveform I(C1) showing that input current flows through capacitor C1 and is then diverted to switching elements M1 and M2 when both switching elements are turned on. Subplot V shows voltage waveform V_(C1)=V_(cap+)−V_(cap−) between nodes V_(cap+) and V_(cap−) and thus, the effective capacitance measured between nodes V_(cap+) and V_(cap−). This effective capacitance includes the contributions of capacitance C1 and switching elements M1 and M2.

In some implementations, the overlap of the gate signals, Vsg1 and Vgs2, can be controlled from zero overlap to complete overlap. When the overlap is zero, all of the input current I₁ flows through capacitor C1 such that the effective capacitance of the PWM capacitor is the value of C1. When the gate signal overlap is complete, all of the input current I₁ flows through the switching elements M1, M2 only. The effective capacitance of the PWM capacitor equals infinity (due to the short circuit effect and thus having an infinitely large capacitance at the frequency of switching). Because the control circuit is able to control the overlap, effective PWM capacitor capacitances from the value of C1 to infinity can be generated.

FIG. 7C shows a zoomed-in view of waveforms of FIG. 7A. Note that subplots I-V in FIG. 7C correspond to zoomed-out views of subplots I-V in FIG. 7B. Window 710 shows that body diode conduction time is greatly reduced.

FIGS. 8A-8F show measurements made from an exemplary embodiment of a mixed signal implementation of the control of a PWM capacitor. The measurements include absolute voltage V_(ab) 802 at the output of the inverter 202 of approximately 500 V/div, input current I₁ 804 of approximately 20 A/div, voltage V_(C1) 806 of approximately 100 V/div at capacitor C1, and voltage measurement V_(gs1) 808 of 10 V/div between gate g1 and reference s. In this embodiment, the power level is maintained approximately between 6 kW and 12 kW. As reference voltage V_(ref) is adjusted, the effective capacitance changes (as indicated by V_(C1)). FIG. 8A shows a V_(ref) of 2.5 V. FIG. 8B shows a V_(ref) of 1.4 V. FIG. 8C shows a V_(ref) of 1 V. FIG. 8D shows a V_(ref) of 0.8 V. FIG. 8E shows a V_(ref) 0.5 V. FIG. 8F shows a V_(ref) of 0.3 V.

Digital Implementation

FIG. 9 shows a diagram of an exemplary embodiment of an example digital implementation of a controller for PWM capacitor. This implementation includes a controller 902, zero-crossing detection stage 904, and a power stage 906. The controller 902 communicates with the zero-crossing detection stage 904, which includes a current sensor 908 that produces a voltage signal for the comparator in the zero-crossing detector 910. The zero-crossing detector 910 provides a zero-crossing signal to the controller 902 to indicate when the current crosses zero (e.g., changes polarity). The zero-crossing detection stage 904 is coupled to power stage 906. The power stage 906 includes signal isolation circuitry 912 for the gate driver 914 input signals. The controller 902 provides the input signals for the gate driver 914. Gate drivers 914 drive switching elements M1 and M2 coupled in parallel with capacitor C1. The current sensors 908 provides a current sense signal to the zero-crossing detector 910. An output of the zero-crossing detector 910 is provided to controller 902 which generates driving signals for transistors M1 and M2. The controller 902 can be implemented as one or more processors or microcontrollers. In some implementations, controller 902 can be implemented as an ASIC or FPGA controller.

In operation, controller 902 controls the effective capacitance of capacitor C1 by alternately switching transistors M1 and M2 in order to bypass or short capacitor C1 for a portion of both the positive and negative half of an AC input voltage signal. An input signal is provided to the controller 902 that indicates a desired effective capacitance for capacitor C1. The controller 902 determines on and off times for the transistors M1 and M2 based on the input signal. In some implementations, the input signal is a phase delay φ ranging between 90 and 180 degrees. The controller 902 determines first and second delay periods from a trigger point of an input current based on the phase delay φ. The controller 902 controls the gate drivers 914 to generate PWM signals for driving the transistors M1 and M2 based on the delay times. For purposes of explanation, the input current zero-crossing is used as a trigger point. However, in some implementations, a current peak can be used as a trigger point. For instance, zero-crossing detector can be modified to detect current peaks by, for example, incorporating a differentiator circuit. In such n implementations, the range for the phase delay φ input may be shifted by 90 degrees to account for the shift in the trigger point.

In general, the controller 902 calculates a transistor turn off delay period and a transistor turn on delay period. The controller 902 receives a zero-crossing signal from the zero-crossing detector 910 and waits for the transistor turn off delay time before turning off the first transistor (e.g., M1). The controller 902 then waits until after the turn on delay period from the zero-crossing to turn the first transistor back on. Another zero-crossing of the current will occur while the first transistor is turned off. In some implementations, the transistor turn on delay period can be measured from the same zero-crossing as the transistor turn off delay period, or, in some implementations, the transistor turn on delay period can be measured from the zero-crossing that occurs while the transistor is turned off. The process is repeated for the second transistor, during the next half cycle of the input current signal.

The transistor turns off and turn on delay times may be the same for both transistors, but triggered from different zero-crossing points (e.g., zero-crossing points occurring at opposite phases of the input current). In some implementations, the turn off and turn and turn on delay times can be different for each transistor. In some implementations, ensuring that the transistors are switched at zero voltage is more critical for turning the transistors on than for turning the transistors off. Therefore, the controller 902 can estimate a theoretical transistor turn on delay based on the phase delay value, as discussed below. In order to ensure that the transistors are turned on when the voltage across capacitor C1 is zero, the controller 902 can wait for an additional period of time after the estimated transistor turn on delay period. In some implementations, the additional period of time is a predetermined delay period (e.g., ≤300 ns, ≤500 ns, ≤800 ns, or ≤1000 ns), for example, to ensure that a body-diode current of a power transistor (or current through an anti-parallel diode) occurs to briefly clamp the voltage across C1 at zero before turning on a transistor. In some implementations, the controller 902 turns the transistor on after the estimated transistor turn on delay period and after detecting body-diode conduction through the transistor (or through an anti-parallel diode). In some implementations, the controller 902 does not estimate a transistor turn on time, but turns on the transistor after detecting body-diode conduction through the transistor (or through an anti-parallel diode). For example, the controller 902 can receive a body-diode conduction signal from a body-diode conduction sensor, such as that discussed in more detail below in reference to FIG. 22 .

FIG. 10A shows a flowchart of an exemplary process 1000 for the control of a PWM capacitor. In some examples, the example process 1000 can be provided as computer-executable instructions executed using one or more processing devices (e.g., processors or microcontrollers) or computing devices. In some examples, the process 1000 may be executed by hardwired electrical circuitry, for example, as an ASIC or an FPGA controller. Process 1000 can be executed by, for example, controller 902.

Step 1002 starts a cycle of a switching period. At step 1004 (time to), the zero-crossing of input current I₁ is detected by the zero-crossing detector 910 when the current I₁ is rising. At step 1006, transistor M1 is scheduled to turn off at time t₂, a turn off delay period after the zero-crossing. For example, a first delay period is calculated based on the input phase φ, where:

$t_{2} = {\frac{\varphi}{360}{{^\circ} \cdot T}}$

and where T is the period of the input current I₁ and the input phase φ sets equivalent capacitance to approximately:

$C_{eq} = {C{1 \cdot {\frac{1}{2 - {\left( {{2\varphi} - {\sin 2\varphi}} \right)/\pi}}.}}}$

At step 1008, transistor M1 is scheduled to turn on at time t₅, a turn on delay period after the zero-crossing and which can be represented by, for example:

$t_{5} = {{\frac{{360^{\circ}} - \varphi}{360{^\circ}} \cdot T} + T_{delay}}$

where predetermined delay T_(delay) is adjusted so zero-voltage switching is ensured. In some implementations, predetermined delay T_(delay) is a fixed delay (e.g., T_(delay)≤300 ns, ≤500 ns, ≤800 ns, or ≤1000 ns). At step 1010 (time t₁), the previous cycle is finished by turning on switching element M2. At step 1012 (time t₂), the transistor M1 is turned off after the turn off delay period. At step 1014 (time t₃), zero-crossing of the input current I₁ is detected when the current is falling. In some implementations, time t₃ is equal to T/2. At step 1016, the transistor M2 is scheduled to turn off at time t₆, a second turn off delay period after the first zero-crossing at to and which can be represented by, for example: t ₆ =T/2+φ/360°·T.

In some implementations, transistor M2 is scheduled to turn off at time t₆ by using the first turn off delay period (calculated above as t₂) but measured from the second zero-crossing of input current I₁ at time t₃.

At step 1018, the transistor M2 is scheduled to turn on at time t₉, a second turn on delay period after the zero-crossing and which can be represented by, for example:

$t_{9} = {{\frac{{480^{\circ}} - \varphi}{360{^\circ}} \cdot T} + {T_{delay}.}}$

In some implementations, transistor M2 is scheduled to turn on at time t₉ by using the first turn on delay period (calculated above as t₅) but measured from the second zero-crossing of input current I₁ at time t₃.

At step 1020 (time t₄), ZVS condition is theoretically achieved for switching element M1 assuming a periodic waveform, such as a sinusoid, for input I₁. In some implementations, time t₄ is estimated by:

$t_{4} = {\frac{{360^{\circ}} - \varphi}{360^{\circ}} \cdot {T.}}$

At step 1022 (time t₅), transistor M1 is turned on after the turn on delay period. At step 1024 (time t₆), transistor M2 is turned off after the second turn off delay period. At step 1026 (time t₇), zero-crossing of input current I₁ is detected to start the next cycle when the current I₁ is rising. Transistor M1 is scheduled to turn off after t=φ/360°·T.

At step 1028 (time t₈), ZVS condition is theoretically achieved for transistor M2 assuming a periodic waveform, such as a sinusoid, for input current I₁. At step 1030 (time t₉), transistor M2 is turned on after the second turn on delay period. Step 1032 is the transition to start the next cycle which leads to step 1012.

FIG. 10B shows a timing diagram of process 1000 described in FIG. 10A. The diagram shows a current I₁ waveform that is marked by vertical lines indicating events. These vertical lines are marked to correspond to steps described in FIG. 10A. Additionally, phase-delay markers 1034, 1036, 1038, 1040 are shown and calculated. At time to, the zero-crossing of rising current I₁ is detected using the zero-crossing detector 910. At time t₁, switching element M2 is switching on (logic 1) and a previous cycle is finished. At time t₂, phase delay 1034 is approximately φ and PWM1 is switched off (logic 0). At time t₃, the zero-crossing of falling current I₁ is detected using zero-crossing detector 910. Time t₄ marks the theoretical M1 body-diode conduction for I₁ current and here phase delay 1036 is approximately 2π−φ. At time t₅, PWM1 is switched on (logic 1) after a delay T_(delay) (between t₄ and t₅) such that ZVS is ensured for all operating conditions. At time t₆, phase delay 1038 is approximately π+φ and PWM2 is switched off (logic 0). At time t₇, the zero-crossing of falling current I₁ is detected using zero-crossing detector 910. Time t₈ marks the theoretical M2 body-diode conduction for sinusoidal I₁ current. At time t₉, PWM1 is switched on after a delay T_(delay) (between t₈ and t₉) such that ZVS is ensured for all operating conditions. Switching on (setting) and switching off (resetting) of signals PWM1 1042 and PWM2 1044 are shown coinciding with time stamps to through t₉.

FIG. 10C shows a flowchart of another exemplary process 1050 for the control of a PWM capacitor. In some examples, the example process 1050 can be provided as computer-executable instructions executed using one or more processing devices (e.g., processors or microcontrollers) or computing devices. In some examples, the process 1050 may be executed by hardwired electrical circuitry, for example, as an ASIC or an FPGA controller. Process 1050 can be executed by, for example, controller 902. Process 1050 is described in reference to the times and events shown in FIG. 10B.

Step 1052 starts a cycle of a switching period. At step 1054 (time to), the controller 902 detects a first zero-crossing of input current I₁, for example, by receiving a zero-crossing detection signal from the zero-crossing detector 910. At step 1056, the controller 902 determines a turn off delay period. For example, the turn of delay period can be determined based on in input value such as an input phase (p. In other words, the input value controls the length of the turn off delay period. For example, the turn off delay can be calculated by: t _(off)=φ/360°·T.

The turn off delay period represents a period of time that the controller waits from each zero-crossing detection until switching off one of the transistors M1 or M2. In some implementations, the turn off delay period determines the effective impedance of the capacitor C1.

At step 1058 (time t₂), the first transistor M1 is turned off after the turn off delay period from the first zero-crossing of the input current I₁. This is represented in FIG. 10B by the PWM1 signal falling to logic zero. At step 1060, the controller 902 measures an elapsed time between switching transistor M1 off and detecting a subsequent (second) zero-crossing of input current I₁ (time t₃). The elapsed time is represented in FIG. 10B by the interval between times t₂ and t₃. For example, the controller 902 can start a counter or timer when transistor M1 is switched off and measure the elapsed time when the next zero-crossing is detected.

At step 1062 (time t₃), the controller 902 detects a second zero-crossing of input current I₁, for example, by receiving a zero-crossing detection signal from the zero-crossing detector 910. At step 1064 controller 902 sets a first turn-on counter based on the elapsed time. For example, the turn-on counter can be set to count down from the elapsed time or the counter that measured the elapsed time can be reversed to count down to zero. The controller 902 uses the turn-on timer to estimate when the voltage across capacitor C1 will return to zero. For instance, as shown in the following FIGS. 11A-11F, the voltage rise and fall across capacitor C1 is genially symmetric about the zero-crossing point of input current I₁. Accordingly, the controller 902 can estimate the theoretical ZVS time (e.g., time t₄) for turning on a transistor (e.g., transistor M1) by counting symmetric times intervals between shutting off the transistor (when the voltage increases in magnitude) and a subsequent zero current crossing (when the voltage reaches a peak) (e.g., t₂-t₃), and between the subsequent zero current crossing and an estimated ZVS time (e.g., t₃-t₄).

At step 1066, the controller 902 turns the first transistor M1 back on after the turn-on counter expires (e.g., after a second delay period measured by the turn-on counter). This is represented in FIG. 10B by the PWM1 signal rising to logic one. Because the turn-on counter is used to estimate a theoretical ZVS time, the controller 902 can incorporate an additional delay T_(delay) before turning on the transistor M1 back on to ensure that zero voltage is achieved. The additional delay T_(delay) is represented in FIG. 10B by the interval between times t₄ and t₅. The additional delay T_(delay) can be a predetermined fixed delay period (e.g., T_(delay) ≤300 ns, ≤500 ns, ≤800 ns, or ≤1000 ns). In some implementations, the additional delay T_(delay) can be a delay between the estimated ZVS time and detecting a zero-voltage condition using a sensor such as a body-diode conduction sensor. For example, the controller 902 can turn the transistor M1 back on in response to a signal from a body-diode conduction sensor (such as that described below in reference to FIG. 22 ). For example, a body-diode conduction sensor can be used to detect detecting body-diode conduction through the transistor (or an associated anti-parallel diode). The controller 902 can use the body-diode conduction as an indication of that a zero voltage condition across the capacitor has been achieved.

At step 1068 (time t₆), the second transistor M2 is turned off after the turn off delay period from the second zero-crossing of the input current I₁ (e.g., at time t₃). This is represented in FIG. 10B by the PWM2 signal falling to logic zero. At step 1070, the controller 902 measures an elapsed time between switching transistor M2 off and detecting a subsequent (third) zero-crossing of input current I₁ (time t₇). The elapsed time is represented in FIG. 10B by the interval between times t₆ and t₇. For example, the controller 902 can start a counter or timer when transistor M2 is switched off and measure the elapsed time when the next zero-crossing is detected.

At step 1072 (time t₇), the controller 902 detects a third zero-crossing of input current I₁, for example, by receiving a zero-crossing detection signal from the zero-crossing detector 910. At step 1074 controller 902 sets a second turn-on counter based on the elapsed time. For example, the second turn-on counter can be set to count down from the elapsed time or the counter that measured the elapsed time can be reversed to count down to zero. The controller 902 uses the turn-on timer to estimate when the voltage across capacitor C1 will return to zero. Accordingly, the controller 902 can estimate the theoretical ZVS time (e.g., time t₈) for turning on a transistor (e.g., transistor M2) by counting symmetric times intervals between shutting off the transistor (when the voltage increases in magnitude) and a subsequent zero current crossing (when the voltage reaches a peak) (e.g., t₆-t₇), and between the subsequent zero current crossing and an estimated ZVS time (e.g., t₇-t₈).

At step 1076, the controller 902 turns the second transistor M2 back on after the second turn-on counter expires (e.g., after a second delay period measured by the turn-on counter). This is represented in FIG. 10B by the PWM2 signal rising to logic one. Because the turn-on counter is used to estimate a theoretical ZVS time, the controller 902 can incorporate an additional delay T_(delay) before turning on the transistor M2 back on to ensure that zero voltage is achieved. The additional delay T_(delay) is represented in FIG. 10B by the interval between times t₈ and t₉. As discussed above, the additional delay T_(delay) can be a predetermined fixed delay period (e.g., T_(delay)≤300 ns, ≤500 ns, ≤800 ns, or ≤1000 ns). In some implementations, the additional delay T_(dday) can be a delay between the estimated ZVS time and detecting a zero-voltage condition using a sensor such as a body-diode conduction sensor. Step 1078 is the transition to start the next cycle which leads to step 1058.

FIGS. 11A-11F show measurements made from an exemplary embodiment of a digital implementation of the control of a PWM capacitor. The measurements include absolute voltage V_(ab) 802 at the output of the inverter 202 of approximately 500 V/div, input current I₁ 804 of approximately 20 A/div, voltage V_(C1) 806 of approximately 100 V/div at capacitor C1, and voltage measurement V_(gs1) 808 10 V/div between gate g1 and reference s. In this embodiment, the power level is maintained approximately between 6 kW and 12 kW. As phase delay φ is adjusted, the effective capacitance changes (as indicated by V_(C1)). FIG. 11A shows a phase φ of 180 degrees. FIG. 11B shows a phase of φ of 140 degrees. FIG. 11C shows a phase φ of 120 degrees. FIG. 11D shows a phase φ of 110 degrees. FIG. 11E shows a phase φ of 100 degrees. FIG. 11F shows a phase φ of 90 degrees.

Protection and Diagnostics

FIG. 12 shows an illustrative mixed-signal implementation 1200 of a PWM controlled capacitor C1 with equivalent capacitance controlled by switching elements M1, M2 and protection/diagnostic functionality. In some implementations, a controller 1202, modulator 1204, and power stage 1206 can have some commonality to the embodiments described above. The power stage 1206 includes a capacitor C1 and switching elements M1, M2 and a current sensor 1208 for sensing current through the capacitor C1. The current sensor 1208 provides capacitor current information CS1, CS2 that can be provided to one or more of a protection/diagnostic circuitry 1210, a peak detector 1212, and a zero-crossing detector 1214. Implementations can include all, none, or any combination of the circuitry receiving the current sensor information CS1, CS2.

The modulator 1204 can includes a reference voltage generator 1217 and a band-pass filter or integrator 1218, which can be similar to that described above. The power stage 1206 can include a signal isolation circuitry 1222 and gate driver 1224, which may be similar to that described above.

FIG. 13A shows an illustrative peak detector 1300 which be provided as the peak detector 1212 of FIG. 12 . The example peak detector 1300 as shown in FIG. 13A uses a bipolar (e.g., +5V and −5V) voltage supply. In the illustrated implementation, the peak detector 1300 includes an op-amp differentiator 1302 and zero-crossing circuit 1304 with low pass filtering and hysteresis. The peak detector 1300 receives capacitor current information CS1, CS2 from current sensor 1208 (FIG. 12 ) and outputs an input current maximum signal CF and minimum signal CR, as shown in FIG. 13B. In some implementations, the rising edge of CF corresponds to an input current maximum and a rising edge of CR corresponds to an input current minimum.

FIG. 13C shows an illustrative peak detector 1300 which be provided as the peak detector 1212 of FIG. 12 . The example peak detector 1300 as shown in FIG. 13C uses a unipolar (e.g., +3.3V) voltage supply. In the illustrated implementation, the peak detector 1300 includes a 1.5V DC bias circuit 1303 and zero-crossing circuit 1304 with low pass filtering and hysteresis. The peak detector 1300 receives capacitor current information CS1 (or CS2) from current sensor 1208 (FIG. 12 ) and outputs an input current maximum signal CF. In some implementations, the rising edge of CF corresponds to an input current maximum. In addition, AC waveforms such as CS1 current measurement are normalized to the +1.5 V dc voltage bias. DC voltage bias can be generated using, for example, resistive dividers, voltage references, shunts and regulators, operational amplifiers, dc-dc converters, or a combination thereof. The slope of the comparators' respective outputs can be controlled by load resistors and capacitors.

FIG. 14A and FIG. 14B show example circuit implementations for current shape OK (CSOK) diagnostics, which is represented in FIG. 12 as CSOK in the protection/diagnostic circuitry 1210. Implementations of a PWM capacitor system can include all, none or any combination of the protection/diagnostic functionality described herein. Furthermore, all, none, or any combination of the protection/diagnostic functionality described herein can be implemented in any combination of hardware and software, including any suitable programmable devices.

The CSOK circuitry checks if input current is “sinusoidal” without discontinuity at zero. In the illustrated embodiment, capacitor current information CS1, CS2 is provided to an op amp that outputs a current information signal CS_SE (FIG. 14A), which is compared to respective positive and negative thresholds (FIG. 14B) and latched by the CF and CR signals. The latch outputs are logically OR'd to provide a CSOKF signal, which is shown in the waveform diagram of FIG. 14C. The CSOK circuit checks on consecutive input current maximums and minimums to determine whether one of them is smaller than a specified threshold value, which can be set to about 0.5-10 A, for example. If any of the consecutive maximums and minimums are larger than respective threshold values, a CSOKF signal is pulled down as an indication that the input current has an acceptable shape.

FIG. 15 shows an illustrative over current protection circuitry which can be seen in FIG. 12 as OCP in the protection/diagnostic circuitry 1210. In the illustrated embodiment, the OCP circuitry uses the CS_SE signal (FIG. 14A), which is provided to respective comparators that check if the input current is above respective positive and negative thresholds OCL+, OCL−. The comparator outputs are logically OR'd and the output is used to latch an error signal to enable a microcontroller to read the error signal (OCEF—over-current error flag).

FIG. 15A shows example waveforms where OCP+ is set to 26 A and OCP− is set to −26 A. As can be seen, subplot I shows the input current, subplot II shows the OECF+ and OECF− signals output from the comparators, and subplot III shows the OCEF signal which is set (latch output) when the input current exceeds about +/−26 A.

FIG. 16 shows an example incremental over-current protection circuit, which is represented as IOCP in the protection/diagnostic circuitry 1210 of FIG. 12 . In some implementations, the IOCP circuit detects large transients where input current is increasing with an exponential envelope. As will be appreciated, such transients are typically caused by faults in the system.

In the illustrated embodiment, the above-described CS_SE is provided to a series of comparators with inputs of maximum and minimum current levels. The comparator outputs are latched with the CF, CR signals and the latch outputs are combined to identify over current conditions.

As shown in the waveform diagram of FIG. 16A, consecutive maximum and minimum current levels are monitored. If a difference in current level between consecutive maximum and minimum levels is greater than a threshold, error signal OCEF_(diff) will be latched until reset by the controller. Subplots I, II, III show an illustrative input current, CF, and CR signal, respectively, as described above. Subplot IV shows an illustrative A signal, which is the value of a comparator output to detect a maximum current level latched in by the CF signal and subplot V shows an illustrative C signal, which is the value of a comparator output to detect a minimum current level latched in by the CR signal. Subplot VI shows an example OCED_(diff) signal which can correspond to a logical AND of the A and C signals.

FIG. 17 shows an example over-voltage protection circuit, which is represented as OVP in the protection/diagnostic circuitry 1210 of FIG. 12 . In general, the OVP circuit uses information from a previous cycle to protect from over-voltage conditions in the current cycle. In some implementations, a premature turn-off of the switching element, such as MOSFET, is prevented by delaying the turn-off edge of a driving PWM signal for a switching element.

FIG. 17A shows example waveforms including the input current, CF signal, delayed CF signal, and voltage zero cross signal, as shown. The PWM_1 is delayed to generate signal C1 that delays turning off switching elements to protect against over voltage conditions in the current cycle.

In some implementations, the protection/diagnostic circuitry 1210 can further include over temperature protection (OTP) having a temperature sensor that can generate an error signal if the measured temperature exceeds a given threshold.

FIG. 18A shows an example implementation of a zero-crossing detector which can correspond to zero-crossing detector 1214 in FIG. 12 . This example implementation of the zero-crossing detector can be a modified or different version of the zero-crossing detector shown in FIG. 5A. The zero-crossing detector can generate differential output signal V_(ZC+), V_(ZC).

FIG. 19 shows an example implementation of a bandpass filter/integrator or ramp generation circuit which can correspond to band-pass filter or integrator 1618 in FIG. 16 . This example implementation of the ramp generation circuit can be a modified or different version of the ramp generator 508 of FIG. 5A. The bandpass filter/integrator can generate a ramp signal, such as the ramp signal shown in FIG. 5B, subplot III.

FIG. 20 shows an example implementation of a PWM signal generator which can correspond to PWM signal generation circuitry 1220 in FIG. 12 . This example implementation of the PWM signal generator can be a modified or different version of the PWM generation circuit in the modulator 404 in FIG. 5A. As described above, the PWM signal generator can generate drive signals for the switching elements, such as M1, M2 in FIG. 12 .

FIG. 21 shows an example digital implementation 2100 including protection/diagnostic functionality that may have some commonality with the system of FIG. 12 . In the illustrated embodiment, a controller 2102 forms part of a modulator 2104, which includes a peak detector 2106 and zero-crossing detector 2108 that may be similar to that shown in conjunction with FIG. 12 . The peak detector 2106 and zero-crossing detector 2108 may receive sensor output signals CS1, CS2 from the power stage 2120. The modulator 2104 may include all, none or any combination of protection/diagnostic circuitry 1210 functionality shown in FIG. 12 and/or FIGS. 13-20 . In the illustrated embodiment, a protection circuitry 2110 can include over voltage protection (OVP) 2112 and over temperature protection (OTP) 2114. In some implementations, the OVP 2112 and OTP 2114 can be similar to the functionality shown and described above in conjunction with FIG. 12 and FIG. 17 , for example. Note that, in some implementations, the microcontroller 2102 can be configured or programmed to perform some or all of the function of the mixed signal implementation. For some functionalities, additional hardware may be required to achieve similar functionality. Functionalities, for example, that can be programmed into the microcontroller 2102 can be the over current protection (OCP), incremental over current protection (iOCP), current shape OK (CSOK), and/or band-pass filter/integrator.

The power stage 2120 can include a signal isolation circuitry 1222 and gate driver 1224, which may be similar to that described above. The power stage 2120 can include a capacitor C1 and switching elements M1, M2 and a current sensor for sensing current through the capacitor C1 and providing current information signals, CS1, CS2, as described above, for example.

Automatic Zero-Voltage Switching Control

In some implementations, a system having a PWM-controlled capacitor includes enhanced circuit for zero-voltage switching of its switches (e.g. MOSFETs).). In some implementations, an automatic ZVS implementation provides ZVS in the presence of relatively significant signal transients to reduce or eliminate switching element, e.g., MOSFET, breakdown relating to the PWM-controlled capacitor. In some implementations, a body diode conduction sensor detects body diode conduction in the switching element and affects switching element control signals, as described more fully below.

FIG. 22 shows an illustrative mixed-signal implementation 2200 of a PWM-controlled capacitor C1 with equivalent capacitance controlled by switching elements M1, M2 and ZVS functionality. In some implementations, a controller 2202, modulator 2204, and power stage 2206 can have some commonality to the embodiments described above. The power stage 2206 includes a capacitor C1 and switching elements M1, M2, which can include internal or external body diodes D1, D2, and a current sensor 2208 for sensing current through the capacitor C1. The current sensor 2208 provides capacitor current information CS1, CS2 that can be provided to a zero-crossing detector 2214, for example, in the modulator 2204.

In some implementations, the power stage 2206 includes a body diode conduction sensor 2215 that can detect conduction of a body diode, e.g., D1, D2, of a switching element, such as M1 or M2 MOSFETS. As described more fully below, a voltage across a sense resistor Rdcs at nodes s1, s2 can be provided to the body diode conduction sensor 2215.

The modulator 2204 can include a reference voltage generator 2218, a band-pass filter or integrator 2220 coupled to the zero-crossing detector 2214, and a PWM signal generator 2222 to generate controls signals for the switching elements M1, M2 which can be similar to those described above. The power stage 2206 can include a signal isolation circuitry 2224 and gate driver 2226, which may be similar to that described above, as well as the body diode conduction sensor 2215. A ZVS circuitry 2230 can be provided between the modulator 2204 and the power stage 2206. In some implementations, the body diode conduction sensor 2215 can be coupled to the controller 2202 via a controller interface 2203.

FIG. 23A shows an example implementation of a zero-crossing detector which can correspond to zero-crossing detector 2214 in FIG. 22 . The zero-crossing detector 2214 receives capacitor current information signals CS1, CS2 as input and generates an output signal CP. The output signal CP is provided to the controller 2202. For example, rising and falling edges of signal CP indicate zero-crossings of the capacitor current. In some implementations, the to zero-crossing detector 2214 can be configured as shown in FIG. 18 and described above.

The example zero-crossing detector 2214 as shown in FIG. 23A uses a unipolar (e.g., +3.3V) voltage supply. In some implementations, the zero-crossing detector 2214 can be configured to use a bipolar (e.g., +5V and −5V) voltage supply (e.g., as shown in FIG. 18 . Furthermore, the comparators can include hysteresis that prevents faulty current pulse detections. In addition, AC waveforms such as current measurement can be normalized to the +1.5 V dc voltage bias. DC voltage bias can be generated using, for example, resistive dividers, voltage references, shunts and regulators, operational amplifiers, dc-dc converters, or a combination thereof. The slope of the comparator outputs can be controlled by load resistors and capacitors.

FIG. 23B shows an example embodiment of the body diode conduction sensor 2215 of FIG. 22 . The example body diode conduction sensor 2215 as shown in FIG. 23A uses a bipolar (e.g., +5V and −5V) voltage supply. In some implementations, the body diode conduction sensor 2215 can be configured to use a unipolar (e.g., 3.3 V) voltage supply. As noted above, the body diode conduction sensor 2215 receives the voltage at nodes s1, s2 at each terminal of the sense resistor Rdcs (FIG. 22 ). In some implementations, the body diode conduction sensor 2215 includes a rail-to-rail comparator 2302 having a first input coupled to node s2 via R7 and a second input coupled to node s1 via R8, with a capacitor C4 coupled across the first and second inputs. The comparator 2302 provides differential outputs Vp, Vn, which are fed back to inputs of the comparator via R9 and R10.

In one embodiment, where switching elements M1, M2 are provided as MOSFETs, when the body-diode for M1, for example, begins to conduct, a current pulse in the sense resistor Rdcs is detected. Components R7, R8, and C4 form a low-pass filter to reduce noise due to ringing of the M1 (or M2) current. Components R7, R8, R9, R10 provide hysteresis for the comparator 2302 that prevents faulty current pulse detections. A rising edge of output V_(n) corresponds to the detection of M1 body-diode start of conduction and a rising edge of output V_(p) corresponds to the detection of M2 body-diode start of conduction. In some implementations, outputs V_(n) and V_(p) are complementary signals.

FIG. 24A-24E show example waveforms for automatic ZVS in accordance with illustrative embodiments of the invention. FIG. 24A shows a waveform for a voltage V(Vcap+, s1) (see FIG. 22 ), which corresponds to the voltage across M1 and FIG. 24B shows a waveform V(Vcap−,s2), which is the voltage across M2. FIG. 24C shows a current I(Rdcs) across the sense resistor Rdcs (see FIG. 22 ). As noted above, when body diode conduction begins, a current pulse is detected across sense resistor Rdcs. In the illustrated embodiment, at time t₁, M2 (see FIG. 24B) begins body diode conduction and the M2 voltage drops to nearly zero as body diode conduction clamps the M2 voltage, which generates a current pulse across the sense resistor Rdcs. At time t₂, M2 is turned on by the M2 gate drive signal (voltage across PWM2, s12 (see FIGS. 22 and 23 )), as shown in FIG. 24E. FIG. 24D shows the gate drive signal for M1. As can be seen, current pulses on Rdcs correspond to body diode conduction of the M1, M2 switching elements. Time td, which is the body diode conduction time, corresponds to t₂-t₁. Furthermore, shorter body diode conduction times, td, can correspond to reduced losses because the voltage drop on the MOSFET channel is lower than the voltage drop of the body diode D1, D2. In some implementations, automatic ZVS waits until detecting M1 or M2 voltage going to zero until enabling the corresponding switch (e.g., M1/M2) to turn on.

FIGS. 25A-25C show further example waveforms for an illustrative automatic ZVS implementation. FIG. 25A shows the voltage across capacitor C1 (V(V_(cap+), V_(cap−)). At time t_(zva), zero voltage across C1 is achieved, which corresponds to a current pulse across the sense resistor Rdcs caused by body diode conduction of M1 or M2, as shown in FIG. 25B. As described above, the current pulse results in a change in the comparator 2302 (FIG. 23 ) output V(V_(n)), which is shown in FIG. 25C.

FIG. 26 shows an example modulator circuit implementation having similarity with the modulator of FIG. 5A. Other circuit implementations are possible that can include programmable devices and various partitions between hardware and software. Operation of the modulator of FIG. 26 is briefly described below.

Resistor R1 converts current signal CS1, CS2 (see FIG. 22 ) from a current to a voltage signal. Components R13, R14, R6, R21, Cf1, Cf2, and comparator CMP form a zero-crossing detector, operation of which is described above. Components R13, R14, Cf1, Cf2, form a low-pass filter to attenuate common-mode and differential mode noise and R13, R14, R6 and R21 provide hysteresis in the zero-crossing detector. Component R2, R4, Cd1, Cd2, R3, R5, C2, C3, Cdcf, Rdcf, and op amp OA includes a band-pass filter/integrator. The integrator function converts a square-wave signal V_(zc)=V_(zc−)−V_(zc+) to a triangular waveform V_(ramp). The band-pass filter function limits the impact of noise. In some implementations V_(ref)=V_(ref+)=−V_(ref), where V_(ref) determines the turn-off edge of the gate driver signals for M1, M2, which can be provided as power MOSFETs, and the equivalent impedance of the PWM capacitor C1. PWM generation circuit outputs PWM_M1 n, PWM_M1, PWM_M2 n, and PWM_M2.

FIGS. 27A-27E shows example waveforms for modulator operation for ZVS. FIG. 27A shows current waveform I(L3), which is current through inductor L3 s of FIG. 2 , for example. FIG. 27B shows the differential output (V_(zc+), V_(az−)) of the zero-crossing detector (see FIG. 26 ) including zero-crossing detection at the high peak and low peak of the triangular ramp signal V(V_(ramp)), shown in FIG. 27C, which is output from the band-pass filter/integrator. FIG. 27D shows the M1 gate drive signal V(pwm_m1) output from the PWM signal generation circuit (FIG. 26 ). FIG. 27E shows the voltage across capacitor C1. As can be seen, M1 is turned off at a time at which the voltage across C1 rises from 0V.

FIGS. 28A and 28B show example pulse shaping and logic conditioning circuit implementations to effect ZVS in illustrative embodiments. FIG. 28A shows a pulse shaping circuit 2800 that receives signals PWM_M1 and PWM_M2 output by the PWM signal generator of FIG. 26 . The pulse shaping circuit 2800 delays the turn on edge of signals PWM_M1 and PWM_M2.

FIG. 28 shows a circuit with a logic AND gate A3 with a first input as an enable signal Ton_cond from the controller, for example, and a second input V_(n), which can be provided by the comparator 2302 of FIG. 23 . As described above, a rising edge of output, V_(n) corresponds to the detection of M1 body-diode start of conduction and rising edge of output V_(p) corresponds to the detection of M2 body-diode start of conduction. The AND A3 output is signal, V_(ne), which is signal V_(n) enabled by the controller. Similarly, AND gate A4 generates enabled signal V_(pe). It will be appreciated that the controller can turn-on or turn-off the PWM capacitor C1 switching elements M1, M2, as well as turn-on or turn-off automatic ZVS functionality.

The M1 pulse-shaped gate drive signal PWM_1_PS and V_(ne) signal are provided as input to logic OR gate A1, which outputs M1 gate drive signal PWM_1. The M2 delayed gate drive signal PWM_2_PS and V_(pe) signal are provided as input to logic OR gate A2, which outputs M2 gate drive signal PWM_2.

Signals PWM_M1 and PWM_M2 are modified to PWM_1_PS and PWM_2_PS so that their rising edge in time-domain waveform comes after the rising edge of V_(n) and V_(p) The rising edge of M1 gate driver signal PWM_1 is determined by rising edge of V_(ne), while the falling edge is determined by PWM_1_PS. The rising edge of the M2 gate driver signal PWM_2 is determined by the rising edge of V_(pe), while the falling edge is determined by PWM_2_PS.

FIG. 29A shows current signal I(L3), FIG. 29B shows voltage level for signal PWM_M1 (input to circuit 2800 of FIG. 28A), FIG. 29C shows the voltage level for PWM_1_PS*5, where the “*5” refers to a scaling factor. FIG. 29D shows the voltage across C1, V(V_(cap+), V_(cap−)). As described above, the M1 gate drive signal PMW_M1 is delayed to delay M1 turn on so that automatic ZVS is allowed to complete.

FIGS. 30A-30F show example waveforms for PWM control of capacitor C1 with automatic ZVS in accordance with example embodiments of the invention. FIG. 30A shows the voltage V(v_(cap+), v_(cap−)) across capacitor C1 and FIG. 30B shows a current pulse I(Rdcs) across sense resistor Rdcs (FIG. 25B), resulting in a transition of signal V_(n) in FIG. 30C (also shown in FIG. 25C), to enable M1 to turn on. The delayed turn-on of M1 is shown in FIG. 30D as V(pwm_1_ps), the generation of which is shown in FIG. 28A. The modulator and pulse shaping (see FIGS. 28A, 28B) determine the pulse edge of V(pwm_1_ps) that turns off M1. FIG. 30E shows the gate drive signal to M1, V(PWM1, s12), as the logical OR of V_(n) and pwm_1_ps, as shown in FIG. 28B. The gate drive signal V(PWM2, s12) for M2 shown in FIG. 30F is generated in a similar manner.

FIG. 31A and FIG. 31B show example results for the illustrative test circuit shown in FIG. 31C, which is similar to the circuit of FIG. 2 . PWM of capacitor C3 s (see FIG. 31C) is performed with automatic ZVS, as described above. FIG. 31A shows, on the left, for silicon MOSFET switching devices for M1 and M2, no automatic ZVS, and, on the right, with automatic ZVS. FIG. 31B shows, on the left, for silicon MOSFET switching devices for M1 and M2, with automatic ZVS, and, on the right, without automatic ZVS for a different voltage reference V_(ref) from that of FIG. 31A.

FIG. 32 shows power loss reduction for SiC MOSFETs for M1 and M2 without (left side of FIG. 32 ) and with (right side of FIG. 32 ) automatic ZVS for the circuit of FIG. 31C. As can be seen, Automatic ZVS brings about 16 W of power loss savings.

FIG. 33 shows thermal imaging of M1 and M2 as Sic MOSFETs providing about a 7.3 C temperature reduction during operation of automatic ZVS (right side of FIG. 33 ) as compared to operating without ZVS functionality (left side of FIG. 33 ).

FIG. 34 shows an exemplary computer 3400 that can perform at least part of the processing described herein. The computer 3400 includes a processor 3402, a volatile memory 3404, a non-volatile memory 3406 (e.g., hard disk), an output device 3407 and graphical user interface (GUI) 3408 (e.g., a mouse, a keyboard, a display, for example). The non-volatile memory 3406 stores computer instructions 3412, an operating system 3416 and data 3418. In one example, the computer instructions 3412 are executed by the processor 3402 out of volatile memory 3404. In one embodiment, an article 3420 includes non-transitory computer-readable instructions.

While the disclosed techniques have been described in connection with certain preferred embodiments, other embodiments will be understood by one of ordinary skill in the art and are intended to fall within the scope of this disclosure. For example, designs, methods, configurations of components, etc. related to transmitting wireless power have been described above along with various specific applications and examples thereof. Those skilled in the art will appreciate where the designs, components, configurations or components described herein can be used in combination, or interchangeably, and that the above description does not limit such interchangeability or combination of components to only that which is described herein.

For illustrative purposes, the foregoing description focuses on the use of devices, components, and methods in high power wireless power transfer applications, e.g., power transfer for charging electric vehicles.

More generally, however, it should be understood that devices that can receive power using the devices, components, and methods disclosed herein can include a wide range of electrical devices, and are not limited to those devices described for illustrative purposes herein. In general, any portable electronic device, such as a cell phone, keyboard, mouse, radio, camera, mobile handset, headset, watch, headphones, dongles, multifunction cards, food and drink accessories, and the like, and any workspace electronic devices such as printers, clocks, lamps, headphones, external drives, projectors, digital photo frames, additional displays, and the like, can receive power wirelessly using the devices, components, and methods disclosed herein. Furthermore, any electrical device, such as electric or hybrid vehicles, motorized wheel chairs, scooters, power tools, and the like, can receive power wirelessly using the devices, components, and methods disclosed herein.

In this disclosure, certain circuit or system components such as capacitors, inductors, resistors, are referred to as circuit “components” or “elements.” The disclosure also refers to series and parallel combinations of these components or elements as elements, networks, topologies, circuits, and the like. More generally, however, where a single component or a specific network of components is described herein, it should be understood that alternative embodiments may include networks for elements, alternative networks, and/or the like.

As used herein, the term “coupled” when referring to circuit or system components is used to describe an appropriate, wired or wireless, direct or indirect, connection between one or more components through which information or signals can be passed from one component to another.

As used herein, the term “direct connection” or “directly connected,” refers to a direct connection between two elements where the elements are connected with no intervening active elements between them. The term “electrically connected” or “electrical connection,” refers to an electrical connection between two elements where the elements are connected such that the elements have a common potential. In addition, a connection between a first component and a terminal of a second component means that there is a path between the first component and the terminal that does not pass through the second component.

Implementations of the subject matter and the operations described in this specification can be realized in digital electronic circuitry, or in computer software, firmware, or hardware, including the structures disclosed in this specification and their structural equivalents, or in combinations of one or more of them. Implementations of the subject matter described in this specification can be realized using one or more computer programs, i.e., one or more modules of computer program instructions, encoded on computer storage medium for execution by, or to control the operation of, data processing apparatus. Alternatively or in addition, the program instructions can be encoded on an artificially generated propagated signal, e.g., a machine-generated electrical, optical, or electromagnetic signal that is generated to encode information for transmission to suitable receiver apparatus for execution by a data processing apparatus. A computer storage medium can be, or be included in, a computer-readable storage device, a computer-readable storage substrate, a random or serial access memory array or device, or a combination of one or more of them. Moreover, while a computer storage medium is not a propagated signal; a computer storage medium can be a source or destination of computer program instructions encoded in an artificially generated propagated signal. The computer storage medium can also be, or be included in, one or more separate physical components or media (e.g., multiple CDs, disks, or other storage devices).

The operations described in this specification can be implemented as operations performed by a data processing apparatus on data stored on one or more computer-readable storage devices or received from other sources.

The term “data processing apparatus” encompasses all kinds of apparatus, devices, and machines for processing data, including by way of example a programmable processor, a computer, a system on a chip, or multiple ones, or combinations, of the foregoing. The apparatus can include special purpose logic circuitry, e.g., an FPGA (field programmable gate array) or an ASIC (application-specific integrated circuit). The apparatus can also include, in addition to hardware, code that creates an execution environment for the computer program in question, e.g., code that constitutes processor firmware, a protocol stack, a database management system, an operating system, a cross-platform runtime environment, a virtual machine, or a combination of one or more of them. The apparatus and execution environment can realize various different computing model infrastructures, such as web services, distributed computing and grid computing infrastructures.

A computer program (also known as a program, software, software application, script, or code) can be written in any form of programming language, including compiled or interpreted languages, declarative or procedural languages, and it can be deployed in any form, including as a stand-alone program or as a module, component, subroutine, object, or other unit suitable for use in a computing environment. A computer program may, but need not, correspond to a file in a file system. A program can be stored in a portion of a file that holds other programs or data (e.g., one or more scripts stored in a markup language document), in a single file dedicated to the program in question, or in multiple coordinated files (e.g., files that store one or more modules, sub-programs, or portions of code). A computer program can be deployed to be executed on one computer or on multiple computers that are located at one site or distributed across multiple sites and interconnected by a communication network.

The processes and logic flows described in this specification can be performed by one or more programmable processors executing one or more computer programs to perform actions by operating on input data and generating output. The processes and logic flows can also be performed by, and apparatus can also be implemented as, special purpose logic circuitry, e.g., an FPGA (field programmable gate array) or an ASIC (application-specific integrated circuit).

Processors suitable for the execution of a computer program include, by way of example, both general and special purpose microprocessors, and any one or more processors of any kind of digital computer. Generally, a processor will receive instructions and data from a read-only memory or a random access memory or both. Elements of a computer can include a processor for performing actions in accordance with instructions and one or more memory devices for storing instructions and data. Generally, a computer will also include, or be operatively coupled to receive data from or transfer data to, or both, one or more mass storage devices for storing data, e.g., magnetic, magneto-optical disks, or optical disks. However, a computer need not have such devices. Moreover, a computer can be embedded in another device, e.g., a wireless power transmitter or receiver or a wirelessly charged or powered device such as a vehicle, a mobile telephone, a personal digital assistant (PDA), a mobile audio or video player, a game console, or a Global Positioning System (GPS) receiver, to name just a few. Devices suitable for storing computer program instructions and data include all forms of non-volatile memory, media and memory devices, including by way of example semiconductor memory devices, e.g., EPROM, EEPROM, and flash memory devices; magnetic disks, e.g., internal hard disks or removable disks; magneto-optical disks; and CD-ROM and DVD-ROM disks. The processor and the memory can be supplemented by, or incorporated in, special purpose logic circuitry.

While this specification contains many specific implementation details, these should not be construed as limitations on the scope of any implementation of the present disclosure or of what may be claimed, but rather as descriptions of features specific to example implementations. Certain features that are described in this specification in the context of separate implementations can also be implemented in combination in a single implementation. Conversely, various features that are described in the context of a single implementation can also be implemented in multiple implementations separately or in any suitable sub-combination. Moreover, although features may be described above as acting in certain combinations and even initially claimed as such, one or more features from a claimed combination can in some cases be excised from the combination, and the claimed combination may be directed to a sub-combination or variation of a sub-combination.

Similarly, while operations are depicted in the drawings in a particular order, this should not be understood as requiring that such operations be performed in the particular order shown or in sequential order, or that all illustrated operations be performed, to achieve desirable results. In certain circumstances, multitasking and parallel processing may be advantageous. Moreover, the separation of various system components in the implementations described above should not be understood as requiring such separation in all implementations, and it should be understood that the described program components and systems can generally be integrated together in a single software product or packaged into multiple software products. 

What is claimed is:
 1. An impedance matching network of a wireless power transmission system, comprising: first and second transistor switching elements having internal body diodes or external antiparallel diodes associated therewith; a pulse width modulation (PWM)-switched capacitor coupled across the first and second switching elements; and a controller coupled to the first and second transistor switching elements and configured to control the first and second switching elements by steering current flow away from body diodes into the channels of the first and second transistor switching elements, wherein the controller comprises zero voltage switching (ZVS) circuitry to control switching to occur when a voltage across the PWM-switched capacitor and the first and second switching elements is near or at zero wherein the controller includes: a microcontroller; and a zero-crossing detection stage having an output sent to the microcontroller, wherein the zero-crossing detection stage includes: a comparator, and a current sensor that produces a voltage signal for the comparator; and a power stage to which the zero-crossing detection stage is coupled, wherein the power stage includes: gate drivers for driving the first and second transistor switching elements, and signal isolation for input signals to the gate drivers generated by the microcontroller.
 2. The impedance matching network of claim 1, wherein the controller is a mixed signal implementation.
 3. The impedance matching network of claim 1, wherein the controller is a digital signal implementation that is configured to perform operations comprising: starting a cycle of a switching period; detecting a zero-crossing of an input current by a zero-crossing detector when the input current is rising; scheduling the first transistor switching element to turn off at time t₂, wherein t₂=φ/360°·T, and T is a period of the input current and phase φ sets an equivalent capacitance of the PWM-switched capacitor to approximately: ${C_{eq} = {C{1 \cdot \frac{1}{2 - {\left( {{2\varphi} - {\sin 2\varphi}} \right)/^{\prime}}}}}};$ scheduling the second transistor switching element to turn on at a time t₅, wherein $t_{5} = {{\frac{{360{^\circ}} - \varphi}{360{^\circ}} \cdot T} + T_{delay}}$ and delay T_(delay) is adjusted so zero-voltage switching is ensured for all operating conditions; finishing the cycle by turning on the second transistor switching element; turning off the first transistor switching element; detecting zero-crossing of the input current when the input current is falling; scheduling the second transistor switching element to turn off at time to, wherein t ₆ =T/2+φ/360°·T; scheduling the second transistor switching element to turn on at time t₉, wherein ${t_{9} = {{\frac{{480^{\circ}} - \varphi}{360{^\circ}} \cdot T} + T_{del}}};$ zero voltage switching first transistor switching element; turning on the first transistor switching element; turning off the second transistor switching element; detecting zero-crossing of the input current to start a next cycle when the input current is rising; scheduling switching element to turn off after t=φ/360·T; zero voltage switching the second transistor switching element; turning on the second transistor switching element; and transitioning to a start of a next cycle.
 4. The impedance matching network claim 1, wherein the first and second transistor switching elements are MOSFET devices.
 5. The impedance matching network of claim 1, wherein the first and second transistor switching elements are gallium nitride (GaN) or silicon carbide (Sid) transistor switching elements.
 6. The impedance matching network of claim 1, wherein the controller is a gate control module for providing a first gate control signal for the first switching element and a second gate control signal for the second switching element, as well as a reference potential for a node between the gates of the first and second switching elements.
 7. The impedance matching network of claim 2, wherein the PWM-switched capacitor provides an equivalent capacitance of ${Ceq} = {C1\frac{1}{2 - {\left( {{2\varphi} - {\sin 2\varphi}} \right)/}}}$ where C1 is an impedance value of the capacitor and φ is a phase delay.
 8. An impedance matching network of a wireless power transmission system, comprising: first and second transistor switching elements having internal body diodes or external antiparallel diodes associated therewith; a pulse width modulation (PWM) switched capacitor coupled across the first and second switching elements; and a controller coupled to the first and second transistor switching elements and configured to control the first and second switching elements by steering current flow away from body diodes into the channels of the first and second transistor switching elements, wherein the controller comprises zero voltage switching (ZVS) circuitry to control switching to occur when a voltage across the PWM-switched capacitor and the first and second switching elements is near or at zero; and the controller is a digital signal implementation that is configured to perform operations comprising: starting a cycle of a switching period; detecting a zero-crossing of an input current by a zero-crossing detector when the input current is rising; scheduling the first transistor switching element to turn off at time t₂, wherein t₂=φ/360°·T, and T is a period of the input current and phase φ sets an equivalent capacitance of the PWM-switched capacitor to approximately: ${C_{eq} = {C{1 \cdot \frac{1}{2 - {\left( {{2\varphi} - {\sin 2\varphi}} \right)/\pi^{\prime}}}}}};$ scheduling the second transistor switching; element to turn on at a time t₅, wherein $t_{5} = {{\frac{{360{^\circ}} - \varphi}{360{^\circ}} \cdot T} + T_{delay}}$ and delay T_(delay) is adjusted so zero-voltage switching is ensured for all operating conditions; finishing the cycle by turning on the second transistor switching element; turning off the first transistor switching element; detecting zero-crossing of the input current when the input current is falling; scheduling the second transistor switching element to turn off at time to, wherein t ₆ =T/2+φ/360°·T; scheduling t_(h)e second transistor switching element to turn on at time t₉, wherein ${t_{9} = {{\frac{{480{^\circ}} - \varphi}{360{^\circ}} \cdot T} + T_{del}}};$ zero voltage switching first transistor switching element; turning on the first transistor switching element; turning off the second transistor switching element; detecting zero-crossing of the input current start a next cycle when the input current is rising; scheduling switching element to turn off after t=φ/360°·T; zero voltage switching the second transistor switching element; turning on the second transistor switching element; and transitioning to a start of a next cycle.
 9. The impedance matching network of claim 8, wherein the turn on time t₅ delay T_(delay) is a predetermined time delay.
 10. The impedance matching network of claim 9, wherein the predetermine time delay is a fixed delay of: 300 nanoseconds (ns) or less, 500 ns or less, 800 ns or less, or 1000 ns or less.
 11. The impedance matching network of claim 8, wherein the controller is configured to measure an elapsed time between when the first transistor switching element is turned off at time t₂ and a next zero-crossing that occurs thereafter at a time t₃, wherein time t₃ occurs prior to time t₅ and at an approximate period of T/2.
 12. The impedance matching network of claim 11, wherein the controller sets a counter that counts down from the elapsed time to determine when the voltage across the PWM-switched capacitor and the first and second switching elements is near or at zero.
 13. The impedance matching network of claim 12, wherein the voltage across the PWM-switched capacitor and the first and second switching elements is near or at zero at a time t₄.
 14. The impedance matching network of claim 13, wherein time t₄ occurs between time t₃ and time t₅ and is approximately equal to $\frac{{360{^\circ}} - \varphi}{360{^\circ}} \cdot {T.}$ 